IoT Sensor Node Gets LoRaWAN Certification

Advantech offers its standardized M2.COM IoT LoRaWAN certified sensor node WISE-1510 with integrated ARM Cortex-M4 processor and LoRa transceiver. The module the  is able to provide multi-interfaces for sensors and I/O control such as UART, I2C, SPI, GPIO, PWM and ADC. The WISE-1510 sensor node is well suited for for smart cities, WISE-1510_3D _S20170602171747agriculture, metering, street lighting and environment monitoring. With power consumption optimization and wide area reception, LoRa  sensors or applications with low data rate requirements can achieve years of battery life and kilometers of long distance connection.

WISE-1510 has has received LoRaWAN certification from the LoRa Alliance. Depending on deployment requirements, developers can select to use Public LoRaWAN network services or build a private LoRa system with WISE-3610 LoRa IoT gateway. Advantech’s WISE-3610  is a Qualcomm ARM Cortex A7 based hardware platform with private LoRa ecosystem solution that can connect up to 500 WISE-1510 sensor node devices. Powered by Advantech’s WISE-PaaS IoT Software Platform, WISE-3610 features automatic cloud connection through its WISE-PaaS/WISE Agent service, manages wireless nodes and data via WSN management APIs, and helps customers streamline their IoT data acquisition development through sensor service APIs, and WSN drivers.

Developers can leverage microprocessors on WISE-1510 to build their own applications. WISE-1510 offers unified software—ARM Mbed OS and SDK for easy development with APIs and related documents. Developers can also find extensive resources from Github such as code review, library integration and free core tools. WISE-1510 also offers worldwide certification which allow developers to leverage their IoT devices anywhere. Using Advantech’s WISE-3610 LoRa IoT Gateway, WISE-1510 can be connected to WISE-  PaaS/RMM or  ARM Mbed Cloud service with IoT communication protocols including LWM2M, CoAP, and MQTT. End-to-end integration assists system integrators to overcome complex challenges and helps them build IoT applications quickly and easily.

WISE-1510 features and specifications:

  • ARM Cortex-M4 core processor
  • Compatible support for public LoRaWAN or private LoRa networks
  • Great for low power/wide range applications
  • Multiple I/O interfaces for sensor and control
  • Supports wide temperatures  -40 °C to 85 °C

Advantech | www.advantech.com

Ultrasonic Sensing MCUs Target Smart Water Meters

Texas Instruments has unveiled a new family of MSP430 microcontrollers with an integrated ultrasonic sensing analog front end that enables smart water meters to deliver higher accuracy and lower power consumption. In addition, TI introduced two new reference designs that make it easier to design modules for adding automated meter reading (AMR) capabilities to existing mechanical water meters. The new MCUs and reference designs support the growing demand for more accurate water meters and remote meter reading to enable efficient water resource management, accurate measurement and timely billing.

New ultrasonic MCUs and new reference designs make both electronic and mechanical water meters smarter (PRNewsfoto/Texas Instruments Incorporated)

New ultrasonic MCUs and new reference designs make both electronic and mechanical water meters smarter.

As part of the ultra-low-power MSP430 MCU portfolio for sensing and measurement, the new MSP430FR6047 MCU family lets developers add more intelligence to flow meters by taking advantage of a complete waveform capture feature and analog-to-digital converter (ADC)-based signal processing. This technique enables more accurate measurement than competitive devices, with precision of 25 ps or better, even at flow rates less than 1 liter per hour. In addition, the integrated MSP430FR6047 devices reduce water meter system component count by 50 percent and power consumption by 25 percent, enabling a meter to operate without having to charge the battery for 10 or more years. The new MCUs also integrate a low-energy accelerator module for advanced signal processing, 256 KB of ferroelectric random access memory (FRAM), a LCD driver and a metering test interface.

The MSP430 Ultrasonic Sensing Design Center offers a comprehensive development ecosystem that allows developers to get to market in months. The design center provides tools for quick development and flexibility for customization, including software libraries, a GUI, evaluation modules with metrology and DSP libraries.

TI’s new Low-Power Water Flow Measurement with Inductive Sensing Reference Design is a compact solution for the electronic measurement of mechanical flow meters with low power consumption for longer battery life. Enabled by the single-chip SimpleLink dual-band CC1350 wireless MCU, this reference design also gives designers the ability to add dual-band wireless communications for AMR networks. Designers can take advantage of the reference design’s small footprint to easily retrofit existing mechanical flow meters, enabling water utilities to add AMR capability while avoiding expensive replacement of deployed meters. The CC1350 wireless MCU consumes only 4 µA while measuring water flow rates, enabling longer product life.

A second new reference design is an ultra-low power solution based on the SimpleLink Sub-1 GHz CC1310 wireless MCU. The Low-Power Wireless M-Bus Communications Module Reference Design uses TI’s wireless M-Bus software stack and supports all wireless M-Bus operating modes in the 868-MHz band. This reference design provides best-in-class power consumption and flexibility to support wireless M-Bus deployments across multiple regions.

Texas Instruments | www.ti.com

USB Data Acq System Features Simple Expansion

DATAQ Instruments has announced the release of its model DI-2108-P USB data acquisition (DAQ) system with 16-bit ADC resolution, programmable gain and ChannelStretch technology. The model DI-2108-P provides eight analog input channels each with 2.5-, 5- and 10-volt unipolar and bi-polar programmable measurement ranges. DATAQ Instruments di2108-product-photo-press-releaseThe DI-2108-P also provides 7 digital ports, each configurable as an input or a switch. Two ports can be programmed as counter and frequency measurement inputs. The instrument’s maximum sampling throughput rate is 160 kHz.

The ChannelStretch feature of the DI-2108-P makes channel expansion as easy as adding another device. Plug a second device into a computer and double the channel count of both analog and digital channels. Using USB hubs, plug up to sixteen devices into a single PC for a maximum count of 128 analog and 112 digital channels. And all of them are acquired synchronously at a maximum sample throughput rate of at least 480 kHz. DI-2108-P software support includes ready-to run WinDaq data acquisition software, .Net class, ActiveX controls and a fully documented communication protocol to deploy the instrument on any platform. The unit is priced at $349.

DATAQ Instruments | www.dataq.com

Getting Started with PSoC MCUs (Part 3)

Data Conversion, Capacitive Sensing and More

In the previous parts of this series, Nishant laid the groundwork for getting up and running with the PSoC. Here he tackles the chip’s more complex features like Data Conversion and CapSense.

By Nishant Mittal
Systems Engineer, Cypress Semiconductor

In the previous two parts of this “Getting started with PSoC” series, I have hopefully provided you with a good base of knowledge about PSoC devices. Here, in this final part it’s time to get more in depth and discuss various data conversion protocols in PSoC and provide some design examples. I’ll also cover interfacing various peripherals with the Photo 1microcontroller. We’ll also get into how to transition from a bare silicon PSoC chip or PSoC development board to using the chip in your project.

Data conversion with PSoC

Data Conversion is an important block in any kind of instrumentation system or Internet of Things implementation. In fact, any application that uses sensors or interfaces to the external environment is an application in which Data Conversion is an integral part of the system. Although digital sensors are available today, the lower costs of analog sensors shouldn’t be overlooked.

 

PSoC Creator has a Data Conversion component that enables designers to code efficiently with less effort. The photo above shows the screenshot of the ADC (analog-to-digital conversion) component in PSoC Creator. The photo above also shows the configuration setting for ADC. First off, we need to set the Channel sampling rate (SPS). Second, we need to set the voltage reference which is necessary to do the comparison of analog signals. Here we use VDDA/2 or VDDA which is 5 V. You can select whether you For web Figure 1want a single-ended ADC or differential ADC by simply clicking the appropriate tab from the component configuration. Clock source needs to be chosen. If the source is chosen to be internal, the PLL from the internals of chip are used—otherwise you’d have to connect an external crystal to the controller using the development kit CY8CKIT-044. Other advanced settings are available for complex programs—but most of those aren’t needed in most intermediate applications.

Read the full article in the September 326 issue of Circuit Cellar

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16-Bit, 1.5-Msps Per Channel Octal Simultaneous Sampling SAR ADC

Linear Technology Corp. recently introduced the LTC2320-16 16-bit, 1.5-Msps per channel, no-latency successive approximation register (SAR) ADC. Featuring eight simultaneously sampling channels supporting a rail-to-rail input common mode range, the LTC2320-16 offers a flexible analog front end that accepts fully differential, unipolar or bipolar analog input signals. It also accepts arbitrary input signals and maintains an 82-dB signal-to-noise ratio (SNR) and high common mode rejection ratio (CMRR) of 102 dB when sampling input signals up to the Nyquist frequency. Linear LTC2320-16

 

The LTC2320-16’s specs, features, and benefits:

  • Wide input bandwidth enables the digitization of input signals up to the Nyquist frequency of 750 kHz
  • 1.5 Msps per channel throughput rate
  • Eight simultaneous sampling channels
  • ±2 LSB INL (typ)
  • Guaranteed 16-bit, no missing codes
  • 8.192 VPP true differential inputs with rail-to-rail common mode
  • 82-dB SNR (typ) at fIN = 500 kHz
  • –90-dB THD (Typ) at fIN = 500kHz
  • Guaranteed operation to 125°C
  • Single 3.3- or 5-V supply
  • Low drift (20 ppm/°C max) 2.048- or 4.096-V internal reference
  • 1.8-to-2.5-V I/O voltages
  • CMOS or LVDS SPI-Compatible Serial I/O
  • Power dissipation 20 mW/Ch (typ, 5-V operation)
  • 52-pin 7 mm × 8 mm QFN package

The LTC2320-16 is available in commercial, industrial, and automotive (–40° to 125°C) temperature grades. Pricing begins at $16.50 each in 1,000-piece quantities. The DC2395A evaluation board for the LTC2320 SAR ADC family is available at www.linear.com/demo.

Source: Linear Technology

Fast 16-bit ADC, Four-Channel 14-bit ADC, & Digital Variable Gain Amp

Texas Instruments launched the ADS54J60, which is the industry’s first 16-bit 1-GSPS ADC and the first to achieve over 70 dBFS signal-to-noise ratio (SNR) at 1-GSPS. Texas instruments also announced the highest-density, four-channel, 14-bit 500-MSPS ADC, the ADS54J54. To optimize the signal chain, TI’s new 4.5-GHz LMH6401 fully differential digital variable gain amplifier (DVGA) offers the widest bandwidth with DC coupling and allows signal acquisition of low and high frequencies without the limitation of baluns used in AC-coupled systems. These ADCs work together with the amplifier to provide the highest performance, lowest power and space savings in defense and aerospace, test and measurement, and communication infrastructure applications.Texas Instruments

All ICs are now sampling. The ADS54J54 costs $500 in 1,000-unit quantities. The ADS54J60 will be available in Q4 2015 for $705 in 1,000-unit quantities. The LMH6401 costs $10.95 in 1,000-unit quantities.

Source: Texas Instruments

New AFEs for Single-Phase Smart Meters & Power Monitoring

Microchip Technology has announced the completion of its MCP391X energy-measurement Analog Front End (AFE) family.  The MCP3919 and MCP3912 integrate three and four channels of 24-bit, delta-sigma ADC, respectively. They have an accuracy of 93.5 dB SINAD, –107-dB THD, and 112-dB SFDR for precise signal acquisition and higher-perforce end products.microchipMCP391Xafe

Microchip also announced two new tools to aid in the development of energy systems using the new AFEs.  The MCP3912 Evaluation Board (part # ADM00499) and MCP3919 Evaluation Board (part # ADM00573) are each available for $129.99.

The MCP3912 and MCP3919 AFEs are both available today for sampling and volume production, with prices starting at $1.84 each in 5,000-unit quantities.  Both AFEs are offered in 28-pin QFN and SSOP packages.

Source: Microchip Technology

Measuring Jitter (EE Tip #132)

Jitter is one of the parameters you should consider when designing a project, especially when it involves planning a high-speed digital system. Moreover, jitter investigation—performed either manually or with the help of proper measurement tools—can provide you with a thorough analysis of your product.

There are at least two ways to measure jitter: cycle-to-cycle and time interval error (TIE).

WHAT IS JITTER?
The following is the generic definition offered by The International Telecommunication Union (ITU) in its G.810 recommendation. “Jitter (timing): The short-term variations of the significant instants of a timing signal from their ideal positions in time (where short-term implies that these variations are of frequency greater than or equal to 10 Hz).”

First, jitter refers to timing signals (e.g., a clock or a digital control signal that must be time-correlated to a given clock). Then you only consider “significant instants” of these signals (i.e., signal-useful transitions from one logical state to the other). These events are supposed to happen at a specific time. Jitter is the difference between this expected time and the actual time when the event occurs (see Figure 1).

Figure 1—Jitter includes all phenomena that result in an unwanted shift in timing of some digital signal transitions in comparison to a supposedly “perfect” signal.

Figure 1—Jitter includes all phenomena that result in an unwanted shift in timing of some digital signal transitions in comparison to a supposedly “perfect” signal.

Last, jitter concerns only short-term variations, meaning fast variations as compared to the signal frequency (in contrast, very slow variations, lower than 10 Hz, are called “wander”).

Clock jitter, for example, is a big concern for A/D conversions. Read my article on fast ADCs (“Playing with High-Speed ADCs,” Circuit Cellar 259, 2012) and you will discover that jitter could quickly jeopardize your expensive, high-end ADC’s signal-to-noise ratio.

CYCLE-TO-CYCLE JITTER
Assume you have a digital signal with transitions that should stay within preset time limits (which are usually calculated based on the receiver’s signal period and timing diagrams, such as setup duration and so forth). You are wondering if it is suffering from any excessive jitter. How do you measure the jitter? First, think about what you actually want to measure: Do you have a single signal (e.g., a clock) that could have jitter in its timing transitions as compared to absolute time? Or, do you have a digital signal that must be time-correlated to an accessible clock that is supposed to be perfect? The measurement methods will be different. For simplicity, I will assume the first scenario: You have a clock signal with rising edges that are supposed to be perfectly stable, and you want to double check it.

My first suggestion is to connect this clock to your best oscilloscope’s input, trigger the oscilloscope on the clock’s rising edge, adjust the time base to get a full period on the screen, and measure the clock edge’s time dispersion of the transition just following the trigger. This method will provide a measurement of the so-called cycle-to-cycle jitter (see Figure 2).

Figure 2—Cycle-to-cycle is the easiest way to measure jitter. You can simply trigger your oscilloscope on a signal transition and measure the dispersion of the following transition’s time.

Figure 2—Cycle-to-cycle is the easiest way to measure jitter. You can simply trigger your oscilloscope on a signal transition and measure the dispersion of the following transition’s time.

If you have a dual time base or a digital oscilloscope with zoom features, you could enlarge the time zone around the clock edge you are interested in for more accurate measurements. I used an old Philips PM5786B pulse generator from my lab to perform the test. I configured the pulse generator to generate a 6.6-MHz square signal and connected it to my Teledyne LeCroy WaveRunner 610Zi oscilloscope. I admit this is high-end equipment (1-GHz bandwidth, 20-GSPS sampling rate and an impressive 32-M word memory when using only two of its four channels), but it enabled me to demonstrate some other interesting things about jitter. I could have used an analog oscilloscope to perform the same measurement, as long as the oscilloscope provided enough bandwidth and a dual time base (e.g., an old Tektronix 7904 oscilloscope or something similar). Nevertheless, the result is shown in Figure 3.

Figure 3—This is the result of a cycle-to-cycle jitter measurement of the PM5786A pulse generator. The bottom curve is a zoom of the rising front just following the trigger. The cycle-to-cycle jitter is the horizontal span of this transition over time, here measured at about 620 ps.

Figure 3—This is the result of a cycle-to-cycle jitter measurement of the PM5786A pulse generator. The bottom curve is a zoom of the rising front just following the trigger. The cycle-to-cycle jitter is the horizontal span of this transition over time, here measured at about 620 ps.

This signal generator’s cycle-to-cycle jitter is clearly visible. I measured it around 620 ps. That’s not much, but it can’t be ignored as compared to the signal’s period, which is 151 ns (i.e., 1/6.6 MHz). In fact, 620 ps is ±0.2% of the clock period. Caution: When you are performing this type of measurement, double check the oscilloscope’s intrinsic jitter as you are measuring the sum of the jitter of the clock and the jitter of the oscilloscope. Here, the latter is far smaller.

TIME INTERVAL ERROR
Cycle-to-cycle is not the only way to measure jitter. In fact, this method is not the one stated by the definition of jitter I presented earlier. Cycle-to-cycle jitter is a measurement of the timing variation from one signal cycle to the next one, not between the signal and its “ideal” version. The jitter measurement closest to that definition is called time interval error (TIE). As its name suggests, this is a measure of a signal’s transitions actual time, as compared to its expected time (see Figure 4).

Figure 4—Time interval error (TIE) is another way to measure jitter. Here, the actual transitions are compared to a reference clock, which is supposed to be “perfect,” providing the TIE. This reference can be either another physical signal or it can be generated using a PLL. The measured signal’s accumulated plot, triggered by the reference clock, also provides the so-called eye diagram.

Figure 4—Time interval error (TIE) is another way to measure jitter. Here, the actual transitions are compared to a reference clock, which is supposed to be “perfect,” providing the TIE. This reference can be either another physical signal or it can be generated using a PLL. The measured signal’s accumulated plot, triggered by the reference clock, also provides the so-called eye diagram.

It’s difficult to know these expected times. If you are lucky, you could have a reference clock elsewhere on your circuit, which would supposedly be “perfect.” In that case, you could use this reference as a trigger source, connect the signal to be measured on the oscilloscope’s input channel, and measure its variation from trigger event to trigger event. This would give you a TIE measurement.

But how do you proceed if you don’t have anything other than your signal to be measured? With my previous example, I wanted to measure the jitter of a lab signal generator’s output, which isn’t correlated to any accessible reference clock. In that case, you could still measure a TIE, but first you would have to generate a “perfect” clock. How can this be accomplished? Generating an “ideal” clock, synchronized with a signal, is a perfect job for a phase-locked loop (PLL). The technique is explained my article, “Are You Locked? A PLL Primer” (Circuit Cellar 209, 2007.) You could design a PLL to lock on your signal frequency and it could be as stable as you want (provided you are willing to pay the expense).

Moreover, this PLL’s bandwidth (which is the bandwidth of its feedback filter) would give you an easy way to zoom in on your jitter of interest. For example, if the PLL bandwidth is 100 Hz, the PLL loop will capture any phase variation slower than 100 Hz. Therefore, you can measure the jitter components faster than this limit. This PLL (often called a carrier recovery circuit) can be either an actual hardware circuit or a software-based implementation.

So, there are at least two ways to measure jitter: Cycle-to-cycle and TIE. (As you may have anticipated, many other measurements exist, but I will limit myself to these two for simplicity.) Are these measurement methods related? Yes, of course, but the relationship is not immediate. If the TIE is not null but remains constant, the cycle-to-cycle jitter is null.  Similarly, if the cycle-to-cycle jitter is constant but not null, the TIE will increase over time. In fact, the TIE is closely linked to the mathematical integral over time of the cycle-to-cycle jitter, but this is a little more complex, as the jitter’s frequency range must be limited.

Editor’s Note: This is an excerpt from an article written by Robert Lacoste, “Analyzing a Case of the Jitters: Tips for Preventing Digital Design Issues,” Circuit Cellar 273, 2013.

Real-Time Processing for PCIe Digitizers

Agilent U5303A PCIe 12bit High-Speed DigitizerThe U5303A digitizer and the U5340A FPGA development kit are recent enhancements to Agilent Technologies’s PCI Express (PCIe) high-speed digitizers. The U5303A and the U5340A FPGA add next-generation real-time peak detection functionalities to the PCIe devices.

The U5303A is a 12-bit PCIe digitizer with programmable on-board processing. It offers high performance in a small footprint, making it an ideal platform for many commercial, industrial, and aerospace and defense embedded systems. A data processing unit (DPU) based on the Xilinx Virtex-6 FPGA is at the heart of the U5303A. The DPU controls the module functionality, data flow, and real-time signal processing. This feature enables data reduction and storage to be carried out at the digitizer level, minimizing transfer volumes and accelerating analysis.

The U5340A FPGA development kit is designed to help companies and researchers protect their IP signal-processing algorithms. The FPGA kit enables integration of an advanced real-time signal processing algorithm within Agilent Technologies’s high-speed digitizers. The U5340A features high-speed medical imaging, analytical time-of-flight, lidar ranging, non-destructive testing, and a direct interface to digitizer hardware elements (e.g., the ADC, clock manager, and memory blocks). The FPGA kit includes a library of building blocks, from basic gates to dual-port RAM; a set of IP cores; and ready-to-use scripts that handle all aspects of the build flow.

Contact Agilent Technologies for pricing.

Agilent Technologies, Inc.
www.agilent.com

Energy-Measurement AFEs

Microchip_MCP3913The MCP3913 and the MCP3914 are Microchip Technology’s next-generation family of energy-measurement analog front ends (AFEs). The AFEs integrate six and eight 24-bit, delta-sigma ADCs, respectively, with 94.5-dB SINAD, –106.5-dB THD, and 112-dB Spurious-Free Dynamic Range (SFDR) for high-accuracy signal acquisition and higher-performing end products.

The MCP3914’s two extra ADCs enable the monitoring of more sensors with one chip, reducing its cost and size. The programmable data rate of up to 125 ksps with low-power modes enables designers to scale down for better power consumption or to use higher data rates for advanced signal analysis (e.g., calculating harmonic content).

The MCP3913 and the MCP3914 improve application performance and provide flexibility to adjust the data rate to optimize each application’s rate of performance vs power consumption. The AFEs feature a CRC-16 checksum and register-map lock, for increased robustness. Both AFEs are offered in 40-pin uQFN packages. The MCP3913 adds a 28-pin SSOP package option.

The MCP3913 and the MCP3914 AFEs cost $3.04 each in 5,000-unit quantities. Microchip Technology also announced the MCP3913 Evaluation Board and the MCP3914 Evaluation Board, two new tools to aid in the development of energy systems using these AFEs. Both evaluation boards cost $99.99.

Microchip Technology, Inc.
www.microchip.com

3-D Integration Impact and Challenges

People want transistors—lots of them. It pretty much doesn’t matter what shape they’re in, how small they are, or how fast they operate. Simply said, the more the merrier. Diversity is also good. The more different the transistors, the more useful and interesting the product. And without any question, the cheaper the transistors, the better. So the issue is, how best to achieve as many diverse transistors at the lowest cost possible.

One approach is more chips. Placing a lot of chips close together on a small board will produce a system with many transistors. Another way is more transistors per chip. Keep on scaling the technology to provide more transistors in one or a few chips.

silicon chipThe third option combines these two approaches. Let’s have many chips with many transistors and end up with a huge number of transistors. However, there is a limit to this approach. It’s well understood that scaling is coming to an end. And placing multiple chips on a board can have a terrible effect on a system’s overall speed and power dissipation.

But there is an elegant and intellectually simple solution. Rather than connecting these chips horizontally across a board, connect them vertically, providing N times more transistors, where N is the number of chips stacked one above another. Such vertical, 3-D integration was first broached by William Shockley, co-inventor of the transistor at Bell Labs in 1947. Shockley described the 3-D integration concept in a 1958 patent, which was followed by Merlin Smith and Emanuel Stern’s 1967 patent outlining how best to produce the holes between layers. We now call these inter-layer holes through silicon vias (TSVs). Technology is still catching up to these 3-D concepts.

Three-dimensional integration offers exciting advantages. For example, the vertical distance between layers is much shorter than the horizontal dimensions across a chip. Three-dimensional circuits, therefore, operate faster and dissipate less power than their 2-D equivalent. A 3-D system is shockingly small, permitting it to fit much more conveniently into a tiny space. Think small portable electronics (e.g., credit cards).

But the most exciting advantage of 3-D integration isn’t the small form factor, higher speed, or lower power; it’s the natural ability to support many disparate technologies and functions as one integrated, heterogeneous system. Even better, each chip layer can be optimized for a particular function and technology, since the individual chips can each be developed in isolation. No more trading off different capabilities to combine disparate technologies on the same chip. Now we can use the absolute best technology for each layer and a completely different and optimized technology for a different layer. This approach enables all kinds of novel applications that until now couldn’t have been conceived or would have been cost-prohibitive.

Imagine placing a microprocessor plane below a MEMS-accelerometer plane below an analog plane (with ADCs) below a temperature sensor, all below a video imager (which has to be at the top to “see”). All of these planes fit together into a tiny (smaller than a fingernail) silicon cube while operating at higher speeds and dissipating lower power.

There are technical issues, including: how to best make the TSVs, how to construct the system architecture to fully exploit the system’s 3-D nature, how to deliver power across these multiple planes, how to synchronize this system to best move data around the cube, how to manage system design complexity, and much more.

Two issues rise to the top. The first is power dissipation (specifically, power density). When many transistors switch at a high rate within a tiny volume, the temperature rises, which can impair performance and reliability. I believe this issue, albeit difficult, is technically solvable and simply will require a lot of good engineering.

The real problem is cost. How do we mature this technology quickly enough to drive the costs down to a point where volume commercial applications are possible? Many companies are close to producing tangible 3-D-based products. Cubes of highly dense memory will likely be the first serious and cost-effective product. Early versions are already available. Three-dimensional integration will soon be here in a serious way with what will be a fascinating assortment of all kinds of exciting new products. You won’t have to wait too long.

Evaluating Oscilloscopes (Part 2)

This is Part 2 of my mini-series on selecting an oscilloscope. Rather than a completely thorough guide, it’s more a “collection of notes” based on my own research. But I hope you find it useful, and it might cover a few areas you hadn’t considered.

Last week I mentioned the differences between PC-based and stand-alone oscilloscopes and discussed the physical probe’s characteristics. This week I’ll be discussing the “core” specifications: analog bandwidth, sample rate, and analog-to-digital converter (ADC) resolution.

Topic 1: Analog Bandwidth
Many useful articles online discuss the analog oscilloscope bandwidth, so I won’t dedicate too much time to it. Briefly, the analog bandwidth is typically measured as the “half-power” or -3 dB point, as shown in Figure 1. Half the power means 1/√2 of the voltage. Assume you put a 10-MHz, 1-V sine wave into your 100-MHz bandwidth oscilloscope. You expect to see a 1-V sine wave on the oscilloscope. As you increase the frequency of the sine wave, you would instead expect to see around 0.707 V when you pass a 100-MHz sine wave. If you want to see this in action, watch my video in which I sweep the input frequency to an oscilloscope through the -3 dB point.

Figure 1: The bandwidth refers to the "half-power" or -3 dB  point. If we drove a sine wave of constant amplitude and increasing frequency into the probe, the -3 dB point would be when the amplitude measured in the scope was 0.707 times the initial amplitude.

Figure 1: The bandwidth refers to the “half-power” or -3 dB point. If we drive a sine wave of constant amplitude and increasing frequency into the probe, the -3 dB point would be when the amplitude measured in the scope is 0.707 times the initial amplitude.

Unfortunately, you are likely to be measuring square waves (e.g., in digital systems) and not sine waves. Square waves contain high-frequency components well beyond the fundamental frequency of the wave. For this reason the “rule of thumb”  is to select an oscilloscope with five times the analog bandwidth of the highest–frequency digital signal you would be measuring. Thus, a 66-MHz clock would require a 330-MHz bandwidth oscilloscope.

If you are interested in more details about bandwidth selection, I encourage you to see one of the many excellent guides. Adafruit has a blog post “Why Oscilloscope Bandwidth Matters” that offers more information, along with links to guides from Agilent Technologies and Tektronix.

If you want to play around yourself, I’ve got a Python script that applies analog filtering to a square wave and plots the results, available here. Figure 2 shows an example of a 50-MHz square wave with 50-MHz, 100-MHz, 250-MHz, and 500-MHz analog bandwidth.

Figure 2: This shows sampling a 50-MHz square wave with 50, 100, 250, and 500-MHz of analog bandwidth.

Figure 2: This shows sampling a 50-MHz square wave with 50, 100, 250, and 500 MHz of analog bandwidth.

Topic 2: Sample Rate
Beyond the analog bandwidth, oscilloscopes also prominently advertise the sample rate. Typically, this is in MS/s (megasamples per second) or GS/s (gigasamples per second). The advertised rate is nearly always the maximum if using a single channel. If you are using both channels on a two-channel oscilloscope that advertises 1 GS/s, typically the maximum rate is actually 500 MS/s for both channels.

So what rate do you need? If you are familiar with the Nyquist criterion, you might simply think you should have a sample rate two times the analog bandwidth. Unfortunately, we tend to work in the time domain (e.g., looking at the oscilloscope screen) and not the frequency domain. So you can’t simply apply that idea. Instead, it’s useful to have a considerably higher sample rate compared to analog bandwidth, say, a five times higher sample rate. To illustrate why, see Figure 3. It shows a 25.3-MHz square wave, which I’ve sampled with an oscilloscope with 50-MHz analog bandwidth. As you would expect, the signal rounds off considerably. However, if I only sample it at 100 MS/s, at first sight the signal is almost unrecognizable! Compare that with the 500 MS/s sample rate, which more clearly looks like a square wave (but rounded off due to analog bandwidth limitation).

Again, these figures both come from my Python script, so they are based purely on “theoretical” limits of sample rate. You can play around with sample rate and bandwidth to get an idea of how a signal might look.

Figure 3

Figure 3: This shows sampling a 25.3-MHz square wave at 100 MS/s results in a signal that looks considerably different than you might expect! Sampling at 500 MS/s results in a much more “proper” looking wave.

Topic 3: Equivalent Time Sampling
Certain oscilloscopes have an equivalent time sampling (ETS) mode, which advertises an insanely fast sample rate. For example, the PicoScope 6000 series, which has a 5 GS/s sample rate, can use ETS mode and achieve 200 GS/s on a single channel, or 50 GS/s on all channels.

The caveat is that this high sample rate is achieved by doing careful phase shifts of the A/D sampling clock to sample “in between” the regular intervals. This requires your input waveform be periodic and very stable, since the waveform will actually be “built up” over a longer time interval.

So what does this mean to you? Luckily, many actual waveforms are periodic, and you might find ETS mode very useful. For example, if you want to measure the phase shift in two clocks through a field-programmable gate array (FPGA), you can do this with ETS. At 50 GS/s, you would have 20 ps resolution on the measurement! In fact, that resolution is so high you could measure the phase difference due to a few centimeters difference in PCB trace.

To demonstrate this, I can show you a few videos. To start with, the simple video below shows moving the probes around while looking at the phase difference.

A more practical demonstration, available in the following video, measures the phase shift of two paths routed through an FPGA.

Finally, if you just want to see a sine wave using ETS you can check out the bandwdith demonstration  I referred to earlier in the this article. The video (see below) includes a portion using ETS mode.

 

Topic 4: ADC Resolution
A less prominently advertised feature of certain oscilloscopes is the ADC bit resolution in the front end. Briefly, the ADC resolution tells you how the analog waveform will get mapped to the digital domain. If you have an 8-bit ADC, this means you have 28 = 256 possible numbers the digital waveform can represent. Say you have a ±5 V range on the oscilloscope—a total span of 10 V. This means the ADC can resolve 10 V / 256 = 39.06 mV difference on the input voltage.

This should tell you one fact about digital oscilloscopes: You should always use the smallest possible range to get the finest granularity. That same 8-bit ADC on a ±1 V range would resolve 7.813 mV. However, what often happens is your signal contains multiple components—say, spiking to 7 V during a load switch, and then settling to 0.5 V. This precludes you from using the smaller range on the input, since you want to capture the amplitude of that 7-V spike.

If, however, you had a 12-bit ADC, that 10 V span (+5 V to -5 V) would be split into 212 = 4,096 numbers, meaning the resolution is now 2.551 mV.  If you had a 16-bit ADC, that 10-V range would give you 216 = 65,536 numbers, meaning you could resolve down to 0.1526 mV. Most of the time, you have to choose between a faster ADC with lower (typically 8-bit) resolution or a slower ADC with higher resolution. The only exception to this I’m aware of is the Pico Technology FlexRes 5000 series devices, which allow you to dynamically switch between 8/12/14/15/16 bits with varying changes to the number of channels and sample rate.

While the typical ADC resolution seems to be 8 bits for most scopes, there are higher-resolution models too. As mentioned, these devices are permanently in high-resolution mode, so you have to decide at purchasing time if you want a very high sample rate, or a very high resolution. For example, Cleverscope has always advertised higher resolutions, and their devices are available in 10, 12, or 14 bits. Cleverscope seems to sell the “digitizer” board separately, giving you some flexibility in upgrading to a higher-resolution ADC. TiePie engineering has devices available from 8–14 bits with various sample rate options. Besides the FlexRes device I mentioned, Pico Technology offers some fixed resolution devices in higher 14-bit resolution. Some of the larger manufacturers also have higher-resolution devices, for example Teledyne LeCroy has its High Resolution Oscilloscope (HRO), which is a fixed 12-bit device.

Note that many devices will advertise either an “effective” or “software enhanced” bit resolution higher than the actual ADC resolution. Be careful with this: software enhancement is done via filtering, and you need to be aware of the possible resulting changes to your measurement bandwidth. Two resources with more details on this mode include the ECN magazine article “How To Get More than 8 Bits from Your 8-bit Scope” and the Teledyne LeCroy application note “Enhanced Resolution.” Remember that a 12-bit, 100-MHz bandwidth oscilloscope is not the same as an 8-bit, 100-MHz bandwidth oscilloscope with resolution enhancement!

Using the oscilloscope’s fast Fourier transform (FFT) mode (normally advertised as the spectrum analyzer mode), we can see the difference a higher-resolution ADC makes. When looking at a waveform on the screen, you may think that you don’t care at all about 14-bit accuracy or something similar. However, if you plan to do measurements such as total harmonic distortion (THD), or otherwise need accurate information about frequency components, having high resolution may be extremely important to achieve a reasonable dynamic range.

As a theoretical example I’m using my script mentioned earlier, which will digitize a perfect sine wave and then display the frequency spectrum. The number of bits in the ADC (e.g., quantization) is adjustable, so the harmonic component is solely due to quantization error. This is shown in Figure 4. If you want to see a version of this using a real instrument, I conduct a similar demonstration in this video.

Certain applications may find the higher bit resolution a necessity. For example, if you are working in high-fidelity audio applications, you won’t be too worried about an extremely high sample rate, but you will need the high resolution.

Figure 4: In the frequency domain, the effect of limited quantization bits is much more apparent. Here a 10-MHz pure sine wave frequency spectrum is taken using a different number of bits during the quantization process.

Figure 4: In the frequency domain, the effect of limited quantization bits is much more apparent. Here a 10-MHz pure sine wave frequency spectrum is taken using a different number of bits during the quantization process. (CLICK TO ZOOM)

Coming Up
This week I’ve taken a look at some of the core specifications. I hope the questions to ask when purchasing an oscilloscope are becoming clearer! Next week, I’ll be looking at the software running the oscilloscope, and details such as remote control, FFT features, digital decoding, and buffer types. The fourth and final week will delve into a few remaining features such as external trigger and clock synchronization and will summarize all the material I’ve covered in this series.

Author’s note: Every reasonable effort has been made to ensure example specifications are accurate. There may, however, be errors or omissions in this article. Please confirm all referenced specifications with the device vendor.

Build an Inexpensive Wireless Water Alarm

The best DIY electrical engineering projects are effective, simple, and inexpensive. Devlin Gualtieri’s design of a wireless water alarm, which he describes in Circuit Cellar’s February issue, meets all those requirements.

Like most homeowners, Gualtieri has discovered water leaks in his northern New Jersey home after the damage has already started.

“In all cases, an early warning about water on the floor would have prevented a lot of the resulting damage,” he says.

You can certainly buy water alarm systems that will alert you to everything from a leak in a well-water storage tank to moisture from a cracked boiler. But they typically work with proprietary and expensive home-alarm systems that also charge a monthly “monitoring” fee.

“As an advocate of free and open-source software, it’s not surprising that I object to such schemes,” Gualtieri says.

In February’s Circuit Cellar magazine, now available for membership download or single-issue purchase, Gualtieri describes his battery-operated water alarm. The system, which includes a number of wireless units that signal a single receiver, includes a wireless receiver, audible alarm, and battery monitor to indicate low power.

Photo 1: An interdigital water detection sensor is shown. Alternate rows are lengths of AWG 22 copper wire, which is either bare or has its insulation removed. The sensor is shown mounted to the bottom of the box containing the water alarm circuitry. I attached it with double-stick foam tape, but silicone adhesive should also work.

Photo 1: An interdigital water detection sensor is shown. Alternate rows are lengths of AWG 22 copper wire, which is either bare or has its insulation removed. The sensor is shown mounted to the bottom of the box containing the water alarm circuitry. I attached it with double-stick foam tape, but silicone adhesive should also work.

Because water conducts electricity, Gualtieri sensors are DIY interdigital electrodes that can lie flat on a surface to detect the first presence of water. And their design couldn’t be easier.

“You can simply wind two parallel coils of 22 AWG wire on a perforated board about 2″ by 4”, he says. (See Photo 1.)

He also shares a number of design “tricks,” including one he used to make his low-battery alert work:

“A battery monitor is an important feature of any battery-powered alarm circuit. The Microchip Technology PIC12F675 microcontroller I used in my alarm circuit has 10-bit ADCs that can be optionally assigned to the I/O pins. However, the problem is that the reference voltage for this conversion comes from the battery itself. As the battery drains from 100% downward, so does the voltage reference, so no voltage change would be registered.

Figure 1: This is the portion of the water alarm circuit used for the battery monitor. The series diodes offer a 1.33-V total  drop, which offers a reference voltage so the ADC can see changes in the battery voltage.

Figure 1: This is the portion of the water alarm circuit used for the battery monitor. The series diodes offer a 1.33-V total drop, which offers a reference voltage so the ADC can see changes in the battery voltage.

“I used a simple mathematical trick to enable battery monitoring. Figure 1 shows a portion of the schematic diagram. As you can see, the analog input pin connects to an output pin, which is at the battery voltage when it’s high through a series connection of four small signal diodes (1N4148). The 1-MΩ resistor in series with the diodes limits their current to a few microamps when the output pin is energized. At such low current, the voltage drop across each diode is about 0.35 V. An actual measurement showed the total voltage drop across the four diodes to be 1.33 V.

“This voltage actually presents a new reference value for my analog conversion. The analog conversion now provides the following digital values:

EQ1Table 1 shows the digital values as a function of battery voltage. The nominal voltage of three alkaline cells is 4.75 V. The nominal voltage of three lithium cells is 5.4 V. The PIC12F675 functions from approximately 2 to 6.5 V, but the wireless transmitter needs as much voltage as possible to generate a reliable signal. I arbitrarily coded the battery alarm at 685, or a little above 4 V. That way, there’s still enough power to energize the wireless transmitter at a useful power level.”

Table 1
Battery Voltage ADC Value
5 751
4.75 737
4.5 721
4.24 704
4 683
3.75 661

 

Gaultieri’s wireless transmitter, utilizing lower-frequency bands, is also straightforward.

Photo 2 shows one of the transmitter modules I used in my system,” he says. “The round device is a surface acoustic wave (SAW) resonator. It just takes a few components to transform this into a low-power transmitter operable over a wide supply voltage range, up to 12 V. The companion receiver module is also shown. My alarm has a 916.5-MHz operating frequency, but 433 MHz is a more popular alarm frequency with many similar modules.”

These transmitter and receiver modules are used in the water alarm. The modules operate at 916.5 MHz, but 433 MHz is a more common alarm frequency with similar modules. The scale is inches.

Photo 2: These transmitter and receiver modules are used in the water alarm. The modules operate at 916.5 MHz, but 433 MHz is a more common alarm frequency with similar modules. The scale is inches.

Gualtieri goes on to describe the alarm circuitry (see Photo 3) and receiver circuit (see Photo 4.)

For more details on this easy and affordable early-warning water alarm, check out the February issue.

Photo 3: This is the water alarm’s interior. The transmitter module with its antenna can be seen in the upper right. The battery holder was harvested from a $1 LED flashlight. The box is 2.25“ × 3.5“, excluding the tabs.

Photo 3: This is the water alarm’s interior. The transmitter module with its antenna can be seen in the upper right. The battery holder was harvested from a $1 LED flashlight. The box is 2.25“ × 3.5“, excluding the tabs.

Photo 4: Here is my receiver circuit. One connector was used to monitor the signal strength voltage during development. The other connector feeds an input on a home alarm system. The short antenna reveals its 916.5-MHz operating frequency. Modules with a 433-MHz frequency will have a longer antenna.

Photo 4: Here is my receiver circuit. One connector was used to monitor the signal strength voltage during development. The other connector feeds an input on a home alarm system. The short antenna reveals its 916.5-MHz operating frequency. Modules with a 433-MHz frequency will have a longer antenna.

 

Client Profile: Pico Technology

Pico Technology
320 North Glenwood Boulevard
Tyler, TX 75702

Contact: sales@picotech.com

Embedded Products/Services: Pico Technology’s PicoScope 5000 series uses reconfigurable ADC technology to offer a choice of resolutions from 8 to 16 bits. For more information, visit www.picotech.com/picoscope5000.html.

PicoProduct information: The new PicoScope 5000 series oscilloscopes have a significantly different architecture. High-resolution ADCs can be applied to the input channels in different series and parallel combinations to boost the sampling rate or the resolution.

In Series mode, the ADCs are interleaved to provide 1 GB/s at 8 bits. In Parallel mode, multiple ADCs are sampled in phase on each channel to increase the resolution and dynamic performance (up to 16 bits).

In addition to their flexible resolution, the oscilloscopes have ultra-deep memory buffers of up to 512 MB to enable long captures at high sampling rates. They also feature standard, advanced software, including serial decoding, mask limit testing, and segmented memory.

The PicoScope 5000 series oscilloscopes are currently available at www.picotech.com.

The two-channel, 60-MHz model with built-function generator costs $1,153. The four-channel, 200-MHz model with built-in arbitrary waveform generator (AWG) costs $2,803. The pricing includes a set of matched probes, all necessary software, and a five-year warranty.

CC280: Analog Communications and Calibration

Are you an analog aficionado? You’re in luck. Two articles, in particular, focus on the November issue’s analog techniques theme. (Look for the issue shortly after mid-October, when it will be available on our website.)

Block Diagram

Data from the base adapter is sent by level shifting the RS-232 or CMOS serial data between 9 and 12 V. A voltage comparator at the remote adapter slices the signal to generate a 0-to-5-V logic signal. The voltage on the signal wire never goes low enough for the 5-V regulator to go out of regulation.

These adapters use a combination of tricks. A single pair of wires carries full-duplex serial data and a small amount of power to a remote device for tasks (e.g., continuous remote data collection and control). The digital signals can be simple on/off signals or more complex signals (e.g., RS-232).

These adapters use a combination of tricks. A single pair of wires carries full-duplex serial data and a small amount of power to a remote device for tasks (e.g., continuous remote data collection and control). The digital signals can be simple on/off signals or more complex signals (e.g., RS-232).

Dick Cappels, a consultant who tinkers with analog and mixed-signal projects, presents a design using a pair of cable adapters and simple analog circuits to enable full-duplex, bidirectional communications and power over more than 100 m of paired wires. Why bother when Power Over Ethernet  (PoE), Bluetooth, and Wi-Fi approaches are available?

“In some applications, using Ethernet is a disadvantage because of the higher costs and greater interface complexity,” Cappels says. “You can use a microcontroller that costs less than a dollar and a few analog parts described in this article to perform remote data gathering and control.”

The base unit including the 5-to-15-V power supply is simple for its functionality. The two eight-pin DIP ICs are a voltage comparator and the switching regulator.

The base unit including the 5-to-15-V power supply is simple for its functionality. The two eight-pin DIP ICs are a voltage comparator and the switching regulator.

Cappels’s need for data channels to monitor his inground water tank inspired his design. Because his local municipality did not always keep the tank filled, he needed to know when it was dry so his pumps wouldn’t run without water and possibly become damaged.
“Besides the mundane application of monitoring a water tank, the system would be excellent for other communication uses,” Cappels says, including computer connection to a home weather station and intrusion-detection systems. Bit rates up to 250 kHz also enable the system to be used in two-way voice communication such as intercoms, he says.

Retired engineer David Cass Tyler became interested in writing his series about calibration while working on a consulting project. “I came to realize that some people don’t really know how to approach the issue of taking an analog-to-digital value to actual engineering units, nor how to correct calibration factors after the fact,” Tyler says

In Part 1 of his article series, Tyler notes: “Digital inputs and digital outputs are pretty simple. They are either on or off. However, for ADCs and DACs to be accurate, they must first be calibrated. This article addresses linear ADCs and DACs.” Part 2, appearing in the December issue, will discuss using polynomial curve fitting to convert nonlinear data to real-world engineering values.

In addition to its analog-themed articles, the November issue includes topics ranging from a DIY solar array tracker’s software to power-capped computer systems.

Editor’s Note: Learn more about Circuit Cellar contributors Dick Cappels and David Cass Tyler by reading their posts about their workspaces and favorite DIY tools.